Systems and methods for automatic determination of state of switches in power converters

ABSTRACT

Systems and methods that automatically detect state of switches in power converters are disclosed. In one aspect, a power switch includes a first switch coupled between a power input node and a first terminal of a load, a second switch coupled between the power input node and a second terminal of the load, first and second current sense devices arranged to transmit first and second signals including at least one of a magnitude and polarity of first and second currents through the first and second switches, respectively, a first driver circuit arranged to transmit first control signals to the first switch based at least in part on a voltage at the power input node and the first signal, and a second driver circuit arranged to transmit second control signals to the second switch based at least in part on the voltage at the power input node and the second signal.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims priority to U.S. patent application Ser. No.17/667,335, for SYSTEMS AND METHODS FOR AUTOMATIC DETERMINATION OF STATEOF SWITCHES IN POWER CONVERTERS,” filed Feb. 8, 2022, which claimspriority to U.S. provisional patent application Ser. No. 63/147,603, for“DYNAMIC ZCD THRESHOLD MODULATION” filed on Feb. 9, 2021, the contentsof all of which are hereby incorporated by reference in their entiretyfor all purposes.

FIELD

The present disclosure relates generally to power conversion circuits,and in particular to power conversion circuits that automatically detectstate of power switches that are used in the power conversion circuit.

BACKGROUND OF THE INVENTION

Electronic devices such as computers, servers and televisions, amongothers, employ one or more electrical power conversion circuits toconvert one form of electrical energy to another. Some electrical powerconversion circuits convert a high DC voltage to a lower DC voltageusing a circuit topology called a half bridge converter. As manyelectronic devices are sensitive to size and efficiency of the powerconversion circuit, new power converters can provide relatively higherefficiency and lower size for the new electronic devices.

SUMMARY

In some embodiments, a circuit is disclosed. The circuit includes afirst switch coupled between a power input node and a first terminal ofa load, a second switch coupled between the power input node and asecond terminal of the load, a first current sense device arranged totransmit a first signal including at least one of a magnitude andpolarity of a first current through the first switch, a second currentsense device arranged to transmit a second signal including at least oneof a magnitude and polarity of a second current through the secondswitch, a first driver circuit arranged to transmit first controlsignals to the first switch based at least in part on a voltage at thepower input node and the first signal, and a second driver circuitarranged to transmit second control signals to the second switch basedat least in part on the voltage at the power input node and the secondsignal.

In some embodiments, the first driver circuit includes a first thresholdgeneration circuit and the second driver circuit includes a secondthreshold generation circuit.

In some embodiments, the first threshold generation circuit is arrangedto generate a first threshold signal based on the voltage at the powerinput node.

In some embodiments, a value of the first threshold signal is based on aduty cycle of a pulse width modulated (PWM) signal received from acontroller.

In some embodiments, the value of the first threshold signal is highwhen the duty cycle of the PWM signal is high.

In some embodiments, the value of the first threshold signal is low whenthe duty cycle of the PWM signal is low.

In some embodiments, the second threshold generation circuit is arrangedto generate a second threshold signal based on the voltage at the powerinput node.

In some embodiments, a value of the second threshold signal is based onthe duty cycle of the PWM signals received from the controller.

In some embodiments, the first threshold generation circuit includes afirst PWM signals receiving circuit, and a first resistor coupled to afirst current mirror circuit, the first current mirror circuit coupledto the first PWM signals receiving circuit.

In some embodiments, a method of operating a circuit is disclosed. Themethod includes switching first and second power switches to transferpower from an AC power input node to a load, receiving control signalsfor controlling operation of the first and second power switches,providing a first switch device including a first current sensorarranged to transmit a first signal indicating a polarity of a firstcurrent flowing through the first power switch, providing a secondswitch device including a second current sensor arranged to transmit asecond signal indicating a polarity of a second current flowing throughthe second power switch, generating a reference voltage based on avoltage of the AC power input node, and transmitting a turn off signalto the first power switch when a voltage of the first signal is higherthan the reference voltage.

In some embodiments, the transmitting the turn off signal to the firstpower switch occurs when the first signal is higher than the referencevoltage and a turn off signal for the first power switch is receivedfrom a controller.

In some embodiments, the reference voltage is a first reference voltageand the turn off signal is first turn off signal, and the method furtherincludes transmitting a second turn off signal to the second powerswitch when a voltage of the second signal is high than a secondreference voltage

In some embodiments, the transmitting the second turn off signal to thesecond power switch occurs when the second signal is higher than thesecond reference voltage and the second turn off signal for the firstpower switch is received from a controller.

In some embodiments, the generating the reference voltage based on thevoltage of the AC power input node includes: receiving pulse widthmodulated (PWM) signals, and generating the reference voltage based on avalue of a resistance of a first resistor and a duty cycle of the PWMsignals.

In some embodiments, a power factor correction (PFC) circuit isdisclosed. The PFC circuit includes a first power switch coupled betweena switch node and a first terminal of a load, a second power switchcoupled between the switch node and a second terminal of the load, afirst current sense device arranged to transmit a first signal includingat least one of a magnitude and polarity of a first current through thefirst power switch, a second current sense device arranged to transmit asecond signal including at least one of a magnitude and polarity of asecond current through the second power switch, a first driver circuitarranged to transmit first control signals to the first power switchbased at least in part on a voltage at a power input node and the firstsignal, a second driver circuit arranged to transmit second controlsignals to the second power switch based at least in part on the voltageat the power input node and the second signal, and a controller arrangedto transmit control signals to the first and second power switches.

In some embodiments, in the PFC circuit the first driver circuitincludes a first threshold generation circuit and the second drivercircuit includes a second threshold generation circuit.

In some embodiments, in the PFC circuit the first threshold generationcircuit is arranged to generate the first threshold signal based on thevoltage at the power input node.

In some embodiments, in the PFC circuit the first and second powerswitches are arranged to selectively connect the switch node to thefirst and second terminals of the load.

In some embodiments, in the PFC circuit the connecting of the switchnode to the first terminal of the load is performed when the switch nodeis substantially at a same voltage as the first terminal of the load.

In some embodiments, in the PFC circuit the connecting of the switchnode to the second terminal of the load is performed when the switchnode is substantially at a same voltage as the second terminal of theload.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic of a power conversion circuit accordingto an embodiment of the disclosure;

FIG. 2 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 ;

FIG. 3 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 ;

FIG. 4 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 ;

FIG. 5 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 ;

FIG. 6 is a simplified schematic illustration of a variable currentthreshold switching switch circuit according to an embodiment of thedisclosure;

FIG. 7 is a simplified schematic illustration of control mode operationof the circuit of FIG. 6 ;

FIG. 8 is a simplified schematic illustration of synchronous modeoperation of the circuit of FIG. 6 ;

FIG. 9 illustrates a flowchart of an exemplary method for determining astate of a power switch, and a power switch turn-off method according toan embodiment of the disclosure; and

FIG. 10 is a simplified schematic illustration of a zero currentreference voltage generator according to an embodiment of thedisclosure.

DETAILED DESCRIPTION OF THE INVENTION

Circuits and related techniques disclosed herein relate generally topower converters. More specifically, circuits, devices and relatedtechniques disclosed herein relate to power conversion circuits thatemploy power switches that automatically can detect state of the powerswitches. In some embodiments, circuits, devices and related techniquesdisclosed herein can be used in totem pole bridgeless power factorcorrection (PFC) circuits to improve their operational efficiency byallowing for zero voltage switching (ZVS) of the power switches. Invarious embodiments, ZVS is achieved by determining whether a powerswitch is operating in a control mode or in a synchronous (sync) mode,and if the power switch is operating in sync mode, the power switch maynot turn off in response to a control signal from a controller. Instead,the power switch may automatically and autonomously determine when toturn itself off in order to optimize an amount of current that can flowinto a power conversion inductor such that the energy in the powerconversion inductor can pull a switch node softly down to ground. Inthis way, hard switching of the switch node can be avoided, therebyincreasing the operational efficiency of the totem pole bridgeless PFC.

In some embodiments, the determination of the mode of operation of apower switch (control or sync mode), and the optimization of the amountof current that may flow into the power conversion inductor canperformed by generating a threshold signal that is based on an inputline voltage, generating a current sense signal based on at least one ofdirection and a magnitude of a current flowing through the power switch.In various embodiments, the threshold signal may be based on adifference between the output voltage of the power converter and theinput line voltage. The current sense signal can be compared to thethreshold signal and if the current sense signal is less than thethreshold signal, the power switch may not turn off in response to acontrol signal from the controller. Instead, the power switch can delayits turn off until the current sense signal exceeds the thresholdsignal.

The threshold signal can be based on an instantaneous value of the inputline voltage when the output voltage of the power converter is at afixed value. By having the threshold signal be based on theinstantaneous value of the input line voltage, an amount of a reversecurrent flow into the power conversion inductor can be optimized inorder to achieve ZVS, resulting in an increased efficiency of the powerconverter. In various embodiments, by having a capability toautomatically and autonomously detect whether the power switch isoperating in control or sync mode, computational loads on the controllercan be reduced resulting in system cost reductions.

In some embodiments, the power switch may turn off immediately if itdetermines that it is in control mode, whereas if the power switchdetermines that it is in sync mode the switch may not turn offimmediately. Instead, it may wait until a current through the powerswitch turns positive by a relatively small amount before it turnsitself off. The relatively small amount of positive current isdetermined by the power switch automatically such that the inductor cancharge up by an optimized amount of energy in order to be able to pullthe switch node down to ground softly, i.e. achieve ZVS, resulting in anincreased operational efficiency of the power converter.

Embodiments of the disclosure can enable a power converter to increaseits operational efficiency while increasing a control loop bandwidth ofthe controller. A determination of an on-time of the synch switch can beperformed autonomously by the power switch itself, thereby freeing thecontroller from performing this task, resulting in an increased controlloop bandwidth because less calculation is performed by the controllerduring each control loop update. This can result in improved performanceof the power converter. Further, a less complex controller may beutilized in the power converter, resulting in reduced system costs.

In some embodiments, the totem pole bridgeless PFC circuits can utilizeone or more gallium nitride (GaN) devices, such as GaN powertransistors. By utilizing GaN devices, embodiments of the presentdisclosure can enable the power converter to operate at relativelyhigher frequencies with relatively higher efficiencies than traditionalsilicon-based circuits, because GaN transistors may have relativelylower reverse recovery charge compared to their silicon counterparts.Various inventive embodiments are described herein, including methods,processes, systems, devices, and the like.

FIG. 1 is a simplified schematic of a power conversion circuit 100receiving an AC power input from an AC source Vac, and providing powerto a load 150 according to an embodiment of the invention.

Power conversion circuit 100 includes controller 110, input in inductorL, high side switch S1, low side switch S2, diode D1, diode D2, andoutput capacitor Co.

Diodes D1 and D2 may be implemented using any current rectifyingstructure, such as pn junction diode or as a diode connected transistor,as understood by those of skill in the art. Other current rectifyingstructures may be used, as understood by those of skill in the art. Insome embodiments, diodes D1 and D2 may be implemented with switcheswhich are actively controlled by controller 110, as understood by thoseof skill in the art.

Output capacitor Co may be implemented using any capacitor structure,such as two metal conductor plates separated by a dielectric or as oneor more transistors having their drain and source terminals electricallyshorted, where the drain/source terminal functions as a first plate ofthe capacitor and the gate of transistor functions as a second plate ofthe capacitor, as understood by those of skill in the art. Othercapacitive structures may be used, as understood by those of skill inthe art.

Controller 110 is configured to receive a reference voltage Vref and theoutput voltage VO of the power conversion circuit 100. As understood bythose of skill in the art, in some embodiments, the controller 110 maybe configured to receive a voltage generated based on the output voltageVO instead of the output voltage VO itself. Controller 110 is alsoconfigured to receive a clock signal CLK. Based on the clock signal CLK,the reference voltage Vref, and the output voltage VO, controller 110 isconfigured to generate control signals for each of high side switch S1and low side switch S2. For example, controller 110 may be configured togenerate pulse width modulation (PWM) signals which control theconductivity states of high side switch S1 and low side switch S2.Controller 110 may be configured to generate the control signals forhigh side switch S1 and low side switch S2.

Each of the high side switch S1 and the low side switch S2 can beconfigured to respond to the control signals received from controller110 by becoming either conductive or nonconductive. In some embodiments,each of the high side switch S1 and the low side switch S2 areconfigured to become either conductive or nonconductive in response tothe control signals and in response to an electrical condition of thehigh side switch S1 or the low side switch S2. For example, either orboth of high side switch S1 and low side switch S2 may be configured toreceive a control signal from the controller 110 and wait for aparticular electrical condition to occur before becoming conductive ornonconductive according to the control signal.

As understood by those of skill in the art, the control signals includeopen control signals configured to cause either high side switch S1 orlow side switch S2 to become nonconductive and close control signalsconfigured to cause either high side switch S1 or low side switch S2 tobecome conductive.

The electrical condition may include, for example, that either high sideswitch S1 or low side switch S2 is conducting current, is not conductingcurrent, or is conducting current specifically in either direction. Insome embodiments, the electrical condition may include that either highside switch S1 or low side switch S2 is conducting a current which isgreater or less than a threshold current.

In some embodiments, the electrical condition may additionally oralternatively include that either high side switch S1 or low side switchS2 has a voltage across its drain and source terminals which is greateror less than a threshold voltage.

In some embodiments, high side switch S1 may be configured to receive anopen control signal from the controller 110 and become nonconductive inresponse to an electrical condition that the current through high sideswitch S1 is less than a threshold current. Accordingly, a delayduration occurs after the high side switch S1 receives the open controlsignal from the controller 110 before becoming nonconductive.

Additionally or alternatively, in some embodiments, low side switch S2may be configured to receive an open control signal from the controller110 and become nonconductive in response an electrical condition thatthe current through low side switch S2 is less than a threshold current.Accordingly, a delay duration occurs after the low side switch S2receives the open control signal from the controller 110 before becomingnonconductive.

In current approaches, a controller may control the on and off times ofthe switches used in the power converter. In the present disclosure, theswitches can automatically and autonomously detect the state ofoperation that they are in, i.e. control mode or sync mode. Further, ifa switch determines it is operating in sync mode, it can turn off itselfindependent of a control signal from the controller commanding theswitch to turn off. Instead, the sync switch can monitor a currentflowing through its drain to source and turn itself off when the currentchanges direction and reaches a magnitude determined by an instantaneousvalue of input line voltage and a value of a resistance of a resistor.

FIG. 2 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 . While operating in the condition illustratedin FIG. 2 , the AC power input signal is positive, high side switch S1is nonconductive, and low side switch S2 is conductive.

Accordingly, as indicated in FIG. 2 , the current flows through powerconversion circuit 100 from the positive terminal of AC source of Vacthrough inductor L, through low side switch S2 in a positive direction,through diode D2, and to the negative terminal of AC source Vac.

While operating in the illustrated condition, in response to receivingan open control signal from controller 110 because the polarity of theAC power input signal is positive, current flows in the indicateddirection, and low side switch S2 immediately or substantiallyimmediately becomes nonconductive, or becomes nonconductive regardlessof the state of an electrical condition of low side switch S2 whichwould cause a delay in other operating conditions. An embodiment of aswitch which can be used as low side switch S2 is discussed below withreference to FIG. 6 .

FIG. 3 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 . While operating in the condition illustratedin FIG. 3 , the AC power input signal is positive, high side switch S1is conductive, and low side switch S2 is nonconductive.

Accordingly, as indicated in FIG. 3 , the current flows through powerconversion circuit 100 from the positive terminal of AC source of Vacthrough inductor L, through high side switch S1 in a negative direction,and to the positive plate of capacitor Co. In addition, current flowsfrom the negative plate of capacitor Co, through diode D2, and to thenegative terminal of AC source Vac.

While operating in the illustrated condition, in response to receivingan open control signal from controller 110 because the polarity of theAC power input signal is positive, high side switch S1 does notimmediately become nonconductive. Instead, high side switch S1 becomesnonconductive after additionally experiencing an electrical conditionthat the positive current flowing through high side switch S1 is greaterthan a threshold. Or, using the convention illustrated in FIG. 2 , highside switch becomes nonconductive after additionally experiencing anelectrical condition that the negative current flowing through high sideswitch S1 is less than a threshold.

FIG. 4 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 . While operating in the condition illustratedin FIG. 4 , the AC power input signal is negative, high side switch S1is conductive, and low side switch S2 is nonconductive.

Accordingly, as indicated in FIG. 4 , the current flows through powerconversion circuit 100 from the negative terminal of AC source of Vacthrough diode D1, through high side switch S1 in a positive direction,through inductor L, and to the positive terminal of AC source Vac.

While operating in the illustrated condition, in response to receivingan open control signal from controller 110, because the polarity of theAC power input signal is negative, current flows in the indicateddirection, and high side switch S1 immediately or substantiallyimmediately becomes nonconductive, or becomes nonconductive regardlessof the state of an electrical condition of high side switch S1 whichwould cause a delay in other operating conditions. An embodiment of aswitch which can be used as high side switch S1 is discussed below withreference to FIG. 6 .

FIG. 5 is a simplified schematic illustration of an operating conditionof the circuit of FIG. 1 . While operating in the condition illustratedin FIG. 5 , the AC power input signal is negative, high side switch S1is nonconductive, and low side switch S2 is conductive.

Accordingly, as indicated in FIG. 5 , the current flows through powerconversion circuit 100 from the negative terminal of AC source of Vacthrough diode D1, and to the positive plate of capacitor Co. Inaddition, current flows from the negative plate of capacitor Co, throughlow side switch S2 in a negative direction, through inductor L, and tothe positive terminal of AC source Vac.

While operating in the illustrated condition, in response to receivingan open control signal from controller 110, because the polarity of theAC power input signal is negative, low side switch S2 does notimmediately become nonconductive. Instead, low side switch S2 becomesnonconductive after additionally experiencing an electrical conditionthat the positive current flowing through low side switch S2 is greaterthan a threshold. Or, using the convention illustrated in FIG. 2 , lowside switch S2 becomes nonconductive after experiencing an electricalcondition that the negative current flowing through low side switch S2is less than a threshold.

FIG. 6 is a simplified schematic illustration of a variable currentthreshold switching switch circuit 600 according to an embodiment of thedisclosure. In switch circuit 600, a threshold value used for detectinga current condition for switching can be variable, and be based on aninput line voltage. In some embodiments, the threshold value can bebased on a difference between a value of the output voltage VO incircuit 100 and the input line voltage. Switch circuit 600 can include adrive circuit 610 and current sensing switch circuit 620. Currentsensing switch circuit 620 can include a first switch 630 and a secondswitch 622. In some embodiments, the first switch 630 may be a powerfield effect transistor (FET) capable of carrying relatively largecurrents, and the second switch 622 may be a sense FET, capable ofsensing a relatively small portion of the total current flowing throughthe current sensing switch circuit 620. A ratio of a size of an activearea of the switch 622 to a size of an active area of the switch 630 canbe less than 1.0. In various embodiments, the first switch 630 and thesecond switch 622 can be integrated on the same die. In someembodiments, the first switch 630 and the second switch 622 can beGaN-based transistors integrated on the same die. Switch circuit 600 maybe used as either or both of high side switch S1 and low side switch S2in power conversion circuit 100 of FIGS. 1-5 .

In some embodiments, switch circuit 600 can be formed in silicon, GaN orany other suitable semiconductor material. In various embodiments, boththe drive circuit 610 and the switch circuit 620 can be formed in asilicon substrate. In some embodiments, both the drive circuit 610 andthe switch circuit 620 can be formed in a GaN substrate. In someembodiments, the drive circuit 610 can be formed in a silicon substratewhile the switch circuit 620 can be formed in a GaN substrate. Invarious embodiments, both drive circuit 610 and switch circuit 620 canbe monolithically integrated onto a single die. In some embodiments, thedrive circuit 610 and the switch circuit 620 can be formed on separateindividual die. In various embodiments, the drive circuit 610 and theswitch circuit 620 can be integrated into one electronic package, forexample, but not limited to, into a quad-flat no-lead (QFN) package, orinto a dual-flat no-leads (DFN) package, into a ball grid array (BGA)package. In some embodiments, the drive circuit 610 and the switchcircuit 620 can be individually packaged into an electronic package.

In some embodiments, the first switch 630 can have a first gate terminal636, a first drain terminal 632 and a first source terminal 634. Thesecond switch 622 can have a second gate terminal 628, a second drainterminal 624 and a second source terminal 626. The first gate terminal636 can be connected to the second gate terminal 628, where both gatesare connected to a GATE_IN terminal, the first drain 632 can beconnected to the second drain terminal 624, where both drains areconnected to a drain terminal DRAIN, and the first source terminal 634can be connected to the second source terminal 626, where both sourcesare connected to a source terminal SOURCE. In various embodiments,

Current sensing switch circuit 620 is selectively conductive between itsdrain terminal DRAIN and its source terminal SOURCE according to a gatecontrol signal at its gate terminal GATE_IN. Current sensing switchcircuit 620 is also configured to generate a current sense signal at itscurrent sense terminal CS. The current sense signal can be generated byhaving a sensing current Isense, which flows through the second switch622, flow through a current sensing device 644. Current sensing device644 can be connected between the terminal CS_IN and the SOURCE terminal.In this way, a voltage can be generated indicating a value of thecurrent being conducted between the drain terminal DRAIN and the sourceterminal SOURCE. In some embodiments, the current sensing device 644 canbe a resistor, while in alternative embodiments the current sensingdevice 644 can be a FET.

Drive circuit 610 can include logic and gate drive circuit 612,comparator 614, and a zero current reference generator 616.

Zero current reference generator 616 may receive a reference voltage atnode ZCD. Zero current reference generator 616 may further receive a PWMindication signal corresponding with the gate control signal of thecurrent sensing switch circuit 620. In some embodiments, the referencevoltage at node ZCD can be generated based on a current sourced by zerocurrent reference generator 616 to a resistor (not shown), such as aresistor which is external to a chip having zero current referencegenerator 616 formed thereon.

Based on the reference voltage at node ZCD and the PWM indicationsignal, zero current reference generator 616 can generate a referencevoltage Vzcd. An embodiment of a zero current reference generator 616 isdiscussed in further detail below in FIG. 10 .

In some embodiments, a PWM indication signal can be generated at thenode 650. In alternative embodiments, zero current reference generator616 may generate a current sense reference voltage that is not based onthe PWM indication signal.

Comparator 614 can receive a current sense signal Vcs from the currentsensing switch circuit 620, and can receive a current sense referencevoltage Vzcd from the zero current reference generator 616. Comparator614 can generate a current detection signal based on the current sensesignal Vcs and the current sense reference voltage Vzcd. In response tothe current sense signal Vcs being greater than the current sensereference voltage Vzcd, comparator 614 can generate a current detectionsignal indicating that a positive current is flowing from the drain tothe source of the current sensing switch circuit 620, and that a valueof the current is greater than a current threshold, or that a negativecurrent is flowing from the drain to the source of the current sensingswitch circuit 620, and that a value of the current is less than thecurrent threshold. In response to the current sense signal Vcs beingless than the current sense reference voltage Vzcd, comparator 614 cangenerate a current detection signal indicating that a positive currentis flowing from the drain to the source of the current sensing switchcircuit 620, and that a value of the current is less than the currentthreshold, or that a negative current is flowing from the drain to thesource of the current sensing switch circuit 620, and that a value ofthe current is greater than the current threshold.

Logic and gate drive circuit 612 receives a PWM signal, for example,from a controller, such as controller 110. Logic and gate drive circuit612 also receives the current detection signal from comparator 614. Inresponse to the PWM signal indicating that the switch circuit 600 is tobe nonconductive, logic and gate drive circuit 612 generates an openoutput signal for current sensing switch circuit 620 only after thecurrent detection signal additionally indicates that the positivecurrent flowing from the drain to the source of the current sensingswitch circuit 620 is greater than a current threshold, or that thenegative current flowing from the drain to the source of the currentsensing switch circuit 620 is less than the current threshold. Inresponse to the PWM signal indicating that the switch circuit 600 is tobe conductive, logic and gate drive circuit 612 generates a conductoutput signal for current sensing switch circuit 620 causing currentsensing switch circuit 620 to be conductive, where the conduct outputsignal is generated regardless of the state of the current detectionsignal.

In some embodiments, drive circuit 610 is integrated on a first die orintegrated circuit chip, and current sensing switch circuit 620 isintegrated on a second die or integrated circuit chip. For example, thefirst die or integrated circuit chip may include a silicon semiconductorsubstrate, and the second die or integrated circuit chip may include agallium nitride (GaN) semiconductor substrate.

In some embodiments, the reference voltage at node ZCD can be used tocause the logic and gate drive circuit 612 to generate a GATE_OUT signalwhich is a delayed version of the signal at the node PWM. For example, avoltage at node ZCD may cause the voltage at node Vzcd to be equal tothe ground voltage. In some embodiments, drive circuit 610 can include acomparator 614, which provides a signal to logic and gate drive circuit612 causing the logic and gate drive circuit 612 to generate a GATE_OUTsignal which is a delayed version of the signal at the node PWM inresponse to the voltage at node ZCD being greater than a referencevoltage.

FIG. 7 is a simplified schematic illustration of operation of thethreshold switching switch circuit 600 of FIG. 6 , where circuit 600operates in control mode.

At time T1, the PWM signal goes high, indicating that the switch circuit600 is to be conductive. In response, logic and gate drive circuit 612generates a high GATE_OUT signal that can be received by the currentsensing switch circuit 620 at GATE_IN node causing current sensingswitch circuit 620 to be conductive. The high GATE_OUT signal isgenerated regardless of the state of the current detection signal.

In response to the current sensing switch circuit 620 being conductive,the drain to source current Idrain of the current sensing switch circuit620 increases. In addition, the current sense signal Vcs generated bycurrent sensing switch circuit 620 also increases. Furthermore, once thecurrent sense signal Vcs is greater than the threshold voltage Vzcdgenerated by zero current reference generator 616, the current detectionsignal indicates that the positive current flowing from the drain to thesource of the current sensing switch circuit 620 is greater than acurrent threshold.

At time T2, the PWM signal goes low, indicating that the switch circuit600 is to be or become nonconductive. In response, logic and gate drivecircuit 612 generates a low GATE_OUT signal for current sensing switchcircuit 620 causing current sensing switch circuit 620 to benonconductive. The high GATE_OUT signal is generated substantiallyimmediately because the current detection signal already indicates thatthe positive current flowing from the drain to the source of the currentsensing switch circuit 620 is greater than a current threshold.

Accordingly, when switch 600 is used as either the high side switch S1or the low side switch S2 of power conversion circuit 100, and operatesas described above with reference to FIG. 7 , the high side switch S1 orthe low side switch S2 behaves as a control FET of the power conversioncircuit 100.

FIG. 8 is a simplified schematic illustration of operation of thethreshold switching switch circuit 600 of FIG. 6 , where circuit 600operates in sync mode.

At time T1, the PWM signal goes high, indicating that the switch circuit600 is to be conductive. In response, logic and gate drive circuit 612generates a high GATE_OUT signal that can be received by the currentsensing switch circuit 620 at GATE_IN node causing current sensingswitch circuit 620 to be conductive. The high GATE_OUT signal isgenerated regardless of the state of the current detection signal.

In response to the current sensing switch circuit 620 being conductive,the drain to source current Idrain of the current sensing switch circuit620 increases from an initial negative value. In addition, the currentsense signal Vcs generated by current sensing switch circuit 620 alsoincreases.

At time T2, the PWM signal goes low, indicating that the switch circuit600 is to be or become nonconductive. Logic and gate drive circuit 612does not generate a low GATE_OUT signal for current sensing switchcircuit 620 causing current sensing switch circuit 620 to benonconductive in response to the PWM signal going low because thecurrent sense signal Vcs is less than the threshold voltage Vzcdgenerated by zero current reference generator 616.

At time T3, the current sense signal Vcs becomes greater than thethreshold voltage Vzcd, and the current detection signal indicates thatthe positive current flowing from the drain to the source of the currentsensing switch circuit 620 is greater than a current threshold. Inresponse to both the low level of the PWM signal and the currentdetection signal, logic and gate drive circuit 612 generates a lowGATE_OUT signal for current sensing switch circuit 620 causing currentsensing switch circuit 620 to be nonconductive.

Accordingly, when switch 600 is used as either the high side switch S1or the low side switch S2 of power conversion circuit 100, and operatesas described above with reference to FIG. 8 , the high side switch S1 orthe low side switch S2 behaves as a sync FET of the power conversioncircuit 100.

Therefore, when instances of switch 600 are used as both the high sideswitch S1 and the low side switch S2 of power conversion circuit 100,each of the high side switch S1 and the low side switch S2 automaticallyoperate as described above with reference to FIG. 7 and FIG. 8 ,alternately, according to the automatic detection of when the positivecurrent flowing from the drain to the source of the current sensingswitch circuit 620 is greater than the current threshold with respect tothe PWM signal going low. Therefore, in these embodiments, each of thehigh side switch S1 and the low side switch S2 can automaticallyfunction or can be controlled to automatically function as a sync FET ofthe power conversion circuit 100 and as a control FET of the powerconversion circuit 100, alternately.

For example, when power conversion circuit 100 operates under thecondition described with reference to FIG. 2 , low side switch S2 mayoperate as a control FET and high side switch S1 may operate as a syncFET. Additionally, when power conversion circuit 100 operates under thecondition described with reference to FIG. 3 , low side switch S2 mayoperate as a control FET and high side switch S1 may operate as a syncFET. Furthermore, when power conversion circuit 100 operates under thecondition described with reference to FIG. 4 , low side switch S2 mayoperate as a sync FET and high side switch S1 may operate as a controlFET, and when power conversion circuit 100 operates under the conditiondescribed with reference to FIG. 5 , low side switch S2 may operate as async FET and high side switch S1 may operate as a control FET.

In some embodiments, based on the value of the input voltage across ACsource Vac, and/or based on the value of the corresponding current sensesignal Vcs, the controller 110 of power conversion circuit 100 canensure that the PWM signal goes low for the switch operating as thecontrol FET when or about when the switch operating as the control FETis to turn off, as illustrated in FIG. 7 . In addition, the controller110 of power conversion circuit 100 may ensure that the PWM signal goeslow for the switch operating as the sync FET before the switch operatingas the sync FET is to turn off, as illustrated in FIG. 8 . For example,the controller 110 of power conversion circuit 100 may ensure that thePWM signal goes low for the switch operating as the sync FET while thecorresponding current sense signal Vcs indicates that drain current isnegative in the switch operating as the sync FET, as illustrated in FIG.8 .

Accordingly, in some embodiments, the pulse widths of the PWM signalsmay be different according to whether the switch turned on by the pulseof the PWM signal is operating as a sync FET or a control FET. Forexample, the pulse widths of the PWM signals turning on a sync FET maybe shorter than the pulse widths of the PWM signals turning on a controlFET. The pulse widths of the PWM signals controlling a FET that isoperating in sync mode may be shorter because a FET that is operating insync mode can automatically and autonomously determine its turn-offtime, i.e. a falling edge of the PWM signal received from a controlleris not determinative for a sync FET. Rather, the sync FET can monitor acurrent flowing through its drain to source and turn off when thecurrent changes direction, i.e. the current flows backward throughinductor L), and reaches a magnitude determined by an instantaneousvalue of a line voltage. In some embodiments, the magnitude of thecurrent may further be determined by a resistor, such as an externalresistor.

In various embodiments, the pulse widths of the PWM signals may be thesame or substantially the same regardless of whether the switch turnedon by the pulse of the PWM signal is operating as a sync FET or acontrol FET, as understood by those of skill in the art.

In some embodiments, each of a series of consecutive PWM signals go lowbefore the corresponding current sense signal Vcs becomes greater thanthe threshold voltage Vzcd for both the high side switch S1 and the lowside switch S2 of power conversion circuit 100. In these embodiments,the high side switch S1 and the low side switch S2 behave as sync FETsof the power conversion circuit 100, as understood by those of skill inthe art.

FIG. 9 illustrates a flowchart of an exemplary method 900 fordetermining a state of a power switch in a converter circuit, and apower switch turn-off method in the according to an embodiment of thedisclosure. At block 910, a turn-off PWM signal is received by a switchcircuit, i.e. the PWM signal goes low. At block 920, the switch cangenerate a threshold signal (Vzcd) that is based on an input linevoltage. This threshold signal can vary with varying input line voltage.For example, in a bridgeless PFC circuit, the input voltage Vac may varycausing a threshold signal Vzcd to vary according to an instantaneousvalue of Vac. When input voltage Vac has relatively high values, forexample 240 V, a value of signal Vzcd can be relatively high. In thisway the amount of energy that is used to bring the switch node Vs downto ground can depend on the input line voltage Vac such that anoptimized amount of current will be allowed to flow into the inductor L.

Determining an optimized amount of energy that is used to bring theswitch node Vs down to ground can enable zero voltage switching (ZVS) inthe power converter, resulting in improved efficiency of the powerconverter. When the input line voltage Vac is at its peak, more currentcan be allowed to flow into the inductor L, such that relatively highercharge in the inductor L can be used to transition the switch node Vsfrom a relatively high voltage, for example 400 V, to ground.Embodiments of the disclosure implement a variable threshold signalgenerator that can allow the switch circuit to determine a turn-offpoint for the current flowing to the inductor L in sync mode. Thevariable threshold signal can depend on the input line voltage. Thus, atrelatively high input voltages Vac the switch circuit can allow anincreased amount of reverse current to flow into the inductor L, and atrelatively low input line voltages the switch circuit can allow areduced amount of reverse current to flow into the inductor L.

At block 930, the switch circuit can generate a current sense signalthat is based on at least one of direction and magnitude of the currentflowing through the switch. The current sense signal can be generated bysampling a relatively small amount of current through the switch circuitand feeding it into a current sensing device 644.

At block 940, a value of the current sense signal can be compared to avalue of the threshold signal. In some embodiments, a comparator circuitmay be utilized to perform this comparison. If the value of the currentsense signal is greater than the value of the threshold signal, theswitch circuit can turn off immediately in response to the PWM signalgoing low at block 950. If the value of the current sense signal is notgreater than the value of the threshold signal, the switch circuit maynot turn off immediately in response to the PWM signal going low atblock 960. The switch circuit may wait until the value of the currentsense signal becomes greater than value of the threshold signal, whereat that point the switch circuit turns off at block 970.

It will be appreciated that method 900 is illustrative and thatvariations and modifications are possible. Steps described as sequentialmay be executed in parallel, order of steps may be varied, and steps maybe modified, combined, added or omitted.

FIG. 10 is a simplified schematic illustration of a zero currentreference voltage generator 1000 according to an embodiment of thedisclosure. Zero current reference voltage generator 1000 may be used aszero current reference generator 616 of FIG. 6 to generate a referencevoltage Vzcd. In some embodiments, a resistor 1080 may be used to set avalue of Vzcd. In some embodiments, resistor 1080 may not integrated onthe same die as the other elements of zero current reference voltagegenerator 1000. Zero current reference voltage generator 1000 caninclude resistor 1020 having a resistance value of R, low-pass filtercircuit 1030, and switches 1040, 1050 and 1060. Switches 1040, 1050 and1060 may be FETs. In various embodiments, switch 1040 may be anN-MOSFET, while switches 1050 and 1060 may be P-MOSFETs. Switches 1050and 1060 can be arranged in a current mirror configuration. Alternativereference voltage generators may be used.

Low-pass filter circuit 1030 may be any low pass filter circuit, asunderstood by those of skill in the art. Low-pass filter circuit 1030can receive a PWM signal, and can generate an output at node 1070corresponding to a duty cycle of the received PWM signal. In someembodiments, such as that illustrated in FIG. 6 , the received PWMsignal can be the GATE_OUT signal at node GATE_OUT. In alternativeembodiments, the received PWM signal may be the PWM indication signal atnode PWM illustrated in FIG. 6 . In some embodiments, a control signalmay be used as the received PWM signal.

In the illustrated embodiment, an external ZCD resistor can be connectedto the switch 1050 that is arranged in a diode connected configuration.A current through switch 1050 and the resistor 1080 may be mirrored togenerate current 1095 (I) through switch 1060 and resistor 1020. Thevoltage at node 1090 (Vzcd) will be equal to the I*R. As pulse widths ofthe PWM signal gets longer, the voltage at node 1070 may increase. Whenthe voltage at node 1070 exceeds a value equal to I*R plus the thresholdvoltage of switch 1040, the voltage at node 1090 may start to increase.In this way, a modulation of Vzcd with PWM duty cycle can be achieved.Further, Vzcd can track the input line voltage because the PWM dutycycle can be a proxy for the input line voltage when the output voltageVO is fixed. In some embodiments, when the output voltage VO is notfixed, the PWM duty cycle can be a proxy for a difference between outputvoltage VO and the input line voltage. Thus, Vzcd can be a function ofthe input line voltage and the output voltage VO. It will be understoodby those skilled in the art having the benefit of this disclosure thatalternative methods to affect a modulation of Vzcd based on the inputline voltage can be used, and that those methods are within the scope ofthe disclosure.

Therefore, in response to the PWM signal having a relatively low dutycycle, zero current reference voltage generator 1000 can generate arelatively low reference voltage value at output node Vzcd. Similarly,in response to the PWM signal having a relatively high duty cycle, zerocurrent reference voltage generator 1000 may generate a relatively highreference voltage value at output node Vzcd.

A beneficial result of the zero current reference voltage generator 1000generating a PWM signal dependent reference voltage value is that thereference voltage value may be modified so as to reduce or eliminate orsubstantially eliminate excessive current used to charge or dischargenode Vs, where node Vs is the connection node between switches S1 andS2.

During times when both switch S1 and switch S2 are off, current throughinductor L can cause a voltage at node Vs to either increase ordecrease, depending on the direction of the current. At least to reducenoise, switching loss, and conduction loss, switch S2 is optimallyturned on when the voltage at node Vs has decreased to be equal to theground voltage. Similarly, at least to reduce noise, switching loss, andconduction loss, switch S1 is optimally turned on once the voltage atnode Vs has increased to be equal to the output voltage at output nodeVO.

Insufficient current through inductor L results in switch S1 or switchS2 turning on before the voltage at node Vs has transitioned to eitherthe optimal ground voltage or output voltage. Excessive current throughinductor L results in switch S1 or switch S2 turning on after thevoltage at node Vs has transitioned beyond either the optimal groundvoltage or output voltage. Accordingly, either insufficient or excessivecurrent through the inductor L is undesirable.

A value of the current through inductor L can be affected by the valueof the input voltage across AC source Vac during the transitions of thevoltage at node Vs. Accordingly, the value of the voltage at node Vswhen either switch S2 or switch S1 subsequently turns on can depend onboth the value of the input voltage and the current through inductor Lwhen either switch S1 or switch S2 is turned off. Therefore, in order toreduce, or eliminate, or substantially eliminate variation in the valueof the voltage at node Vs when either switch S2 or switch S1subsequently turns on, the relative time when either switch S1 or switchS2 is turned off may be varied to adjust the current through inductor Lwhen either switch S1 or switch S2 is turned off.

For example, if the value of the input voltage is relatively high, therelative time when either switch S1 or switch S2 is turned off may bedelayed to allow the current through inductor L when either switch S1 orswitch S2 is turned off to increase. Similarly, if the value of theinput voltage is relatively low, the relative time when either switch S1or switch S2 is turned off may be caused to occur earlier to reduce thecurrent through inductor L when either switch S1 or switch S2 is turnedoff.

In some embodiments, controller 110 can vary the duty cycle of the PWMsignal such that the duty cycle of the PWM signal corresponds with thevalue of the voltage input across AC source Vac. Accordingly, the dutycycle of the PWM signal can be used to influence the reference voltagevalue at node Vzcd such that the reference voltage value at node Vzcdvaries so as modify the current through inductor L when either switch S1or switch S2 is turned off to reduce, eliminate, or substantiallyeliminate variation in the duration of the voltage transitions at nodeVs caused by variation in input voltage.

A beneficial result of the zero current reference voltage generator 1000generating a reference voltage value which is dependent on the dutycycle of the PWM signal is that the reference voltage value may bemodified so as to reduce or eliminate or substantially eliminatevariation in the value of the voltage at node Vs when either switch S2or switch S1 turns on.

In some embodiments, controller 110 can vary the duty cycle of the PWMsignal such that the duty cycle of the PWM signal corresponds with thevalue of the voltage input across AC source Vac. Accordingly, in someembodiments, the duty cycle of the PWM signal can be used to influencethe reference voltage value at node Vzcd such that the reference voltagevalue at node Vzcd varies so as to eliminate or substantially eliminatevariation in the value of the voltage at node Vs when either switch S2or switch S1 turns on. As a result, energy occurring in the outputvoltage at node VO at the frequency of the input voltage across ACsource Vac is reduced or eliminated, or substantially eliminated.

In addition, because a reference voltage value for Vzcd can be based onI*R, the reference voltage value at output node Vzcd may have a minimumvalue determined by the current of the current through switch 1060 andthe resistor 1020.

Accordingly, in the illustrated embodiment, the reference voltage valuegenerated at output node Vzcd can be modified according to changes inthe duty cycle of the PWM signal. In alternative embodiments, areference voltage value is generated at output node Vzcd which does notchange with changes in the duty cycle of the PWM signal. For example, inalternative embodiments, a reference voltage generator may generate afixed or substantially fixed reference voltage at output node Vzcd.

Embodiments of the present disclosure can enable the power switches todetermine their turn-off time independent of the controller 110 when thepower switches are operating in sync mode. This is in contrast to thecurrent approaches where a controller may control the on and off timesof the power switches. This can put a relatively high computational loadon the controller and increase system costs in the current approaches.

Embodiments of the present disclosure, can enable efficient operation ofthe power converter. The sync switch can stay on beyond a point where acurrent in the power conversion inductor may reach zero value, such thatsome reverse current can build up in the power conversion inductor. Inthis way, when the sync power switch turns off and there is some energybuilt up in the power conversion inductor, the switch node can softlytransition from a relatively high voltage to a relatively low voltage,thereby allowing the control power switch to turn on with zero rsubstantially zero voltage across it. This allows for zero voltageswitching (ZVS) operation, and can improve the operation efficiency ofthe power convert.

In some embodiments, an amount of energy built up in the powerconversion inductor may be a function of the instantaneous value of theinput line voltage. In various embodiments, the amount of energy builtup in the power conversion inductor may be a function of a differencebetween a value of the output voltage and the instantaneous value of theinput line voltage. In some embodiments, the amount of energy built upin the power conversion inductor can be optimized such that theoperational efficiency of the power converter in increased. An optimumamount of the amount of energy built up in the power conversion inductormay be determined by an optimum amount of current that flows through thesync power switch to the power conversion inductor. In variousembodiments, the power switch can determine this optimum amount ofcurrent autonomously by turning itself off when the current changesdirection and when the current reaches a magnitude determined by theinstantaneous line voltage.

In some embodiments, the automatic and autonomous determination of aturn-off time of the sync power switch can reduce computational loads onthe controller 110 resulting in increased control loop bandwidth andreduced system costs. The controller 110 can get the informationregarding the sync mode of operation based on a polarity of the input ACline. The controller 110 may then send a relatively short PWM pulse tothe sync power switch. The power switch may turn on when it receives aleading edge of the PWM pulse from the controller 110. When a trailingedge of the PWM pulse is received from the controller 110, the powerswitch can detect that it is in sync mode and may not turn itself off.The power switch can perform this detection by checking a direction ofthe current flowing from its drain to its source. If this current isnegative, the power switch may ignore the trailing edge of the PWMpulse. The power switch can monitor the current flowing from its drainto its source and may turn off when the current changes direction, i.e.the current flows back into the power conversion inductor, and reaches avalue that is proportional to the instantaneous input line voltage. Invarious embodiments, this value can be proportional to the instantaneousinput line voltage and a resistor, such as, but not limited to, anexternal resistor.

Once the sync power switch turns off, the switch node may softlytransition to ground using an energy stored in the power conversioninductor. The sync power switch may send an indication signal to thecontroller 110 such that after the switch node has reached a value ofzero or substantially zero volt, the controller 110 may know that thesync power switch is off and that it is safe to turn on the controlpower switch and start the next switching cycle.

As described in details above, embodiments of the present disclosure canincrease a control loop bandwidth of the power converter resulting inimproved performance of the power converter. Further, a relativelysimple controller may be used in the power converter thus reducingsystem costs.

While various embodiments of present invention have been described, itwill be apparent to those of skill in the art that many more embodimentsand implementations are possible that are within the scope of thisinvention. Accordingly, the present invention is not to be restrictedexcept in light of the attached claims and their equivalents.

Though the present invention is disclosed by way of specific embodimentsas described above, those embodiments are not intended to limit thepresent invention. Based on the methods and the technical aspectsdisclosed above, variations and changes may be made to the presentedembodiments by those skilled in the art without departing from thespirit and the scope of the present invention.

What is claimed is:
 1. A circuit comprising: a first circuit including:a first switch coupled between a power input node and a first terminalof a load; a first current sense device arranged to transmit a firstsignal including at least one of a magnitude and polarity of a firstcurrent through the first switch; and a first driver circuit arranged totransmit first control signals to the first switch based at least inpart on a voltage at the power input node and the first signal; and asecond circuit including:  a second switch coupled between the powerinput node and a second terminal of the load; a second current sensedevice arranged to transmit a second signal including at least one of amagnitude and polarity of a second current through the second switch;and a second driver circuit arranged to transmit second control signalsto the second switch based at least in part on the voltage at the powerinput node and the second signal.
 2. The circuit of claim 1, wherein thefirst switch is a gallium nitride (GaN) based switch.
 3. The circuit ofclaim 1, wherein the second switch is a gallium nitride (GaN) basedswitch.
 4. The circuit of claim 1, wherein the first driver circuitcomprises a first threshold generation circuit and the second drivercircuit comprises a second threshold generation circuit.
 5. The circuitof claim 4, wherein the first threshold generation circuit is arrangedto generate a first threshold signal based on the voltage at the powerinput node.
 6. The circuit of claim 5, wherein a value of the firstthreshold signal is based on a duty cycle of a pulse width modulated(PWM) signal received from a controller.
 7. The circuit of claim 6,wherein the value of the first threshold signal is high when the dutycycle of the PWM signal is high.
 8. The circuit of claim 6, wherein thevalue of the first threshold signal is low when the duty cycle of thePWM signal is low.
 9. The circuit of claim 5, wherein the first drivercircuit further comprises a first comparator arranged receive the firstthreshold signal, and wherein the first comparator is further arrangedto compare the first signal to the first threshold signal and generate afirst current detection signal.
 10. The circuit of claim 1, wherein thesecond driver circuit further comprises a second comparator arrangedreceive a second threshold signal, and wherein the second comparator isfurther arranged to compare the second signal to the second thresholdsignal and generate a second current detection signal.
 11. The circuitof claim 4, wherein the first threshold generation circuit comprises: afirst transistor having a first gate terminal, a first source terminaland a first drain terminal, wherein the first gate terminal is coupledto a first receiving circuit; a first current mirror having a firstinput port and a first output port, and arranged to replicate a firstcurrent of the first input port to the first output port; and a firstresistor coupled to the first source terminal and to the first outputport.
 12. The circuit of claim 11, wherein the first receiving circuitcomprises a low-pass filter, and wherein the first drain terminal iscoupled to a DC power supply.
 13. A method of operating a circuit, themethod comprising: providing a first circuit including: a first powerswitch; a current sensor arranged to transmit a first signal indicatingat least one of a magnitude and polarity of a first current through thefirst power switch; and a first driver circuit arranged to transmitfirst control signals to the first power switch based at least in parton a voltage at a power input node and the first signal; providing asecond circuit including: a second power switch; a second current sensorarranged to transmit a second signal indicating at least one of amagnitude and polarity of a second current through the second powerswitch; and a second driver circuit arranged to transmit second controlsignals to the second power switch based at least in part on the voltageat the power input node and the second signal; receiving pulse widthmodulated (PWM) control signals for controlling operation of the firstand second power switches; generating a first reference voltage based onthe voltage of the power input node; and transmitting a first turn-offsignal to the first power switch.
 14. The method of claim 13, whereintransmitting the first turn-off signal to the first power switch occurswhen the first signal is higher than the first reference voltage andwhen a first controller turn-off signal is received by the first powerswitch.
 15. The method of claim 14, further comprising transmitting asecond turn-off signal to the second power switch when the second signalis high than a second reference voltage and when a second controllerturn-off signal is received by the second power switch.
 16. The methodof claim 13, wherein the generating the first reference voltage based onthe voltage of the power input node comprises: receiving PWM controlsignals; and generating the first reference voltage based on a value ofa resistance of a first resistor and a duty cycle of the PWM controlsignals.
 17. The method of claim 16, wherein the generating the firstreference voltage based on the voltage of the power input node furthercomprises: sending the received PWM control signals through a low-passfilter and generating a first voltage corresponding to the duty cycle ofthe received PWM control signals; applying the first voltage to a gateterminal of a first transistor to modulate a first current through thefirst transistor; sending the modulated first current through the firsttransistor to the first resistor; and summing, by the first resistor,the modulated first current through the first resistor with a currentfrom a first current mirror.
 18. The method of claim 17, furthercomprising selectively connecting the power input node to a firstterminal of a load and to a second terminal of the load.
 19. The methodof claim 18, wherein the connecting of the power input node to the firstterminal of the load is performed when a drain terminal of the firstpower switch is approximately equal to a voltage at the first terminalof the load.
 20. The method of claim 19, wherein the connecting of thepower input node to the second terminal of the load is performed when asource terminal of the second power switch is approximately equal to avoltage at the second terminal of the load.